System and method for antenna selection

ABSTRACT

Systems and methods that provide antenna selection in multi-antenna-element communication systems are provided. In one embodiment, a system that selects N antenna elements in an M-antenna-element transmitter or an M-antenna-element receiver, in which N is less than M, may include, for example, M antenna elements in the M-antenna-element transmitter or the M-antenna-element receiver; N RF chains; and a switch coupled to the N RF chains. The M-antenna-element receiver may determine a bit error rate for each possible N antenna element subset of the M antenna elements. The M-antenna-element receiver may determine the particular N antenna element subset with a lowest bit error rate. In response to the determination of the particular N antenna element subset with the lowest bit error rate, the switch may couple the N RF chains to the particular N antenna element subset with the lowest bit error rate.

CROSS-REFERENCE TO RELATED APPLICATIONS

The present application is a CONTINUATION of U.S. application Ser. No.10/957,398, filed Oct. 1, 2004. Said U.S. application Ser. No.10/957,398 makes reference to, claims priority to and claims benefitfrom U.S. Application No. 60/507,843, filed Oct. 1, 2003, entitled“Antenna Selection Method in Multi-Antenna Communication Systems BasedOn Minimum Bit Error Rate”. The above-identified applications are herebyincorporated herein by reference in their entirety.

FEDERALLY SPONSORED RESEARCH OR DEVELOPMENT

[Not Applicable]

MICROFICHE/COPYRIGHT REFERENCE

[Not Applicable]

BACKGROUND OF THE INVENTION

Most current wireless communication systems are composed of nodesconfigured with a single transmit and receive antenna. However, for awide range of wireless communication systems, it has been predicted thatthe performance, including capacity, may be substantially improvedthrough the use of multiple transmit and/or multiple receive antennas.Such configurations form the basis of “smart” antenna techniques. Smartantenna techniques, coupled with space-time signal processing, can beutilized both to combat the deleterious effects of multipath fading of adesired incoming signal and to suppress interfering signals. In this wayboth the performance and capacity of digital wireless systems inexistence or being deployed (e.g., CDMA-based systems, TDMA-basedsystems, WLAN systems, and OFDM-based systems such as IEEE 802.11a/g)maybe improved.

At least some of the impairments to the performance of wireless systemsof the type described above may be at least partially ameliorated byusing multi-element antenna systems designed to introduce a diversitygain and suppress interference within the signal reception process. Thishas been described, for example, in “The Impact of Antenna Diversity Onthe Capacity of Wireless Communication Systems”, by J. H. Winters etal., IEEE Transactions on Communications, vol. 42, no. 2/3/4, pages1740-1751, February 1994. Such diversity gains improve systemperformance by mitigating multipath for more uniform coverage,increasing received signal-to-noise ratio for greater range or reducedrequired transmit power, and providing more robustness againstinterference or permitting greater frequency reuse for higher capacity.

Within communication systems incorporating multi-antenna receivers, aset of M receive antennas may be capable of nulling up to M-1interferers. Accordingly, N signals may be simultaneously transmitted inthe same bandwidth using N transmit antennas, with the transmittedsignal then being separated into N respective signals by way of a set ofN antennas deployed at the receiver. Systems of this type are generallyreferred to as multiple-input-multiple-output (MIMO) systems, and havebeen studied extensively. See, for example, “Optimum combining forindoor radio systems with multiple users”, by J. H. Winters, IEEETransactions on Communications, vol. COM-35, no. 11, November 1987;“Capacity of Multi-Antenna Array Systems In Indoor WirelessEnvironment”, by C. Chuah et al., Proceedings of Globecom '98 Sydney,Australia, IEEE 1998, pages 1894-1899 November 1998; and “FadingCorrelation and Its Effect on the Capacity of Multi-Element AntennaSystems” by D. Shiu et al., IEEE Transactions on Communications, vol.48, no. 3, pages 502-513, March 2000.

Some multi-element antenna arrangements (e.g., some MIMOs) providesystem capacity enhancements that can be achieved using theabove-referenced configurations. Under the assumption of perfectestimates of the applicable channel at the receiver, in a MIMO systemthe received signal decomposes to M “spatially-multiplexed” independentchannels. This results in an M-fold capacity increase relative tosingle-antenna systems. For a fixed overall transmitted power, thecapacity offered by MIMOs scales linearly with the number of antennaelements. Specifically, it has been shown that with N transmit and Nreceive antennas an N-fold increase in the data rate over a singleantenna system can be achieved without any increase in the totalbandwidth or total transmit power. See, e.g., “On Limits of WirelessCommunications in a Fading Environment When Using Multiple Antennas”, byG. J. Foschini et al., Wireless Personal Communications, Kluwer AcademicPublishers, vol. 6, no. 3, pages 311-335, March 1998. In experimentalMIMO systems predicated upon N-fold spatial multiplexing, more than Nantennas are often deployed at a given transmitter or receiver. This isbecause each additional antenna adds to the diversity gain and antennagain and interference suppression applicable to all Nspatially-multiplexed signals. See, e.g., “Simplified processing forhigh spectral efficiency wireless communication employing multi-elementarrays”, by G. J. Foschini et al., IEEE Journal on Selected Areas inCommunications, vol. 17, issue 11, November 1999, pages 1841 -1852.

Although increasing the number of transmit and/or receive antennasenhances various aspects of the performance of MIMO systems, theprovision of a separate RF chain for each transmit and receive antennaincreases costs. Each RF chain is generally comprised a low noiseamplifier, filter, downconverter, and analog-to-digital converter (A/D),with the latter three devices typically being responsible for most ofthe cost of the RF chain. In certain existing single-antenna wirelessreceivers, the single required RF chain may account for in excess of 30%of the receiver's total cost. It is thus apparent that as the number oftransmit and receive antennas increases, overall system cost and powerconsumption may dramatically increase.

Some attempts to address these shortcomings may be found, for example,in United States Patent Publication No. 20020102950 entitled “Method andapparatus for selection and use of optimal antennas in wirelesssystems”; “Capacity of MIMO systems with antenna selection”, by A.Molisch et al., Proceedings of IEEE ICC, Helsinki, Finland, June 2001,vol. 2, pp. 570-574; and “On optimum MIMO with antenna selection”, by R.S. Blum et al., IEEE Communications Letters, vol. 6, issue 8, August2002, pages 322-324, in which a subset of transmit/receive antennas areselected from a larger number of antennas. Since with N-fold spatialmultiplexing, at least N RF chains must be used, typically N out of Mtotal antennas would be chosen at the receiver and/or N out of n_(T)total antennas would be chosen at the transmitter, with M>N and n_(T)>N.

The performance of a system with antenna selection depends, for example,on the criteria used in the selection process. Different criteria usedeven under the same channel condition may well result in a differentselected subset of antennas, thereby yielding different performances.Some of the above-reference documents advocate the maximum capacitycriterion to select the antenna subset. However, capacity is anidealized quantity that may be an unachievable bound because it maynecessitate perfect coding and/or equalization and/or continuousmodulation. In practice, the equalizer is not ideal, limited coding (oreven no coding) and quantized modulation is used.

BRIEF SUMMARY OF THE INVENTION

Some aspects of the present invention may relate to systems and methodsthat provide antenna selection in multi-antenna-element communicationsystems. Some embodiments according to some aspects of the presentinvention may select antenna-element subsets in multi-antenna-elementtransmitters and/or multi-antenna-element receivers.

In one embodiment according to some aspects of the present invention, asystem that selects N antenna elements in an M-antenna-elementtransmitter or an M-antenna-element receiver, in which N is less than M,may include, for example, M antenna elements in the M-antenna-elementtransmitter or the M-antenna-element receiver; N RF chains; and a switchcoupled to the N RF chains. The M-antenna-element receiver may determinea bit error rate for each possible N antenna element subset of the Mantenna elements. The M-antenna-element receiver may determine theparticular N antenna element subset with a lowest bit error rate. Inresponse to the determination of the particular N antenna element subsetwith the lowest bit error rate, the switch may couple the N RF chains tothe particular N antenna element subset with the lowest bit error rate.

In another embodiment according to some aspects of the presentinvention, a method that selects N antenna elements in anM-antenna-element transmitter or an M-antenna-element receiver, in whichN is less than M, may include, for example, one or more of thefollowing: determining a bit error rate for each possible N antennaelement subset of a set of M-antenna elements of the M-antenna elementtransmitter or the M-antenna-element receiver; determining theparticular N antenna element subset with a lowest bit error rate; and,in response thereto, coupling N RF chains of the M-antenna-elementtransmitter or the M-antenna-element receiver to the particular Nantenna element subset with the lowest bit error rate.

In yet another embodiment according to some aspects of the presentinvention, in a receiver having one or more RF chains and a firstplurality of receive antennas, a method that selects a subset of thefirst plurality of receive antennas to receive a transmitted RF signal,may include, for example, one or more of the following: establishing asecond plurality of possible subsets of the first plurality of receiveantennas; determining a second plurality of output bit error rates ofthe receiver corresponding to the second plurality of possible subsetsof the first plurality of receive antennas; identifying a minimum of thesecond plurality of output bit error rates; selecting one of the secondplurality of possible subsets of the first plurality of receive antennascorresponding to the minimum of the second plurality of output bit errorrates; and connecting the one or more RE chains to the receive antennasof the selected one of the second plurality of possible subsets.

In yet another embodiment according to some aspects of the presentinvention, in a transmitter having one or more RF chains and a firstplurality of transmit antennas, a method that selects a subset of thefirst plurality of transmit antennas for transmission of an RF signal asa corresponding plurality of RF output signals subsequently received bya receiver, may include, for example, one or more of the following:establishing a second plurality of possible subsets of the firstplurality of transmit antennas; determining a second plurality of outputbit error rates of the receiver corresponding to the second plurality ofpossible subsets of the first plurality of transmit antennas;identifying a minimum of the second plurality of output bit error rates;selecting one of the second plurality of possible subsets of the firstplurality of transmit antennas corresponding to the minimum of thesecond plurality of output bit error rates; and connecting the one ormore RF chains to the transmit antennas of the selected one of thesecond plurality of possible subsets.

In yet another embodiment according to some aspects of the presentinvention, in a communication system including a transmitter having aplurality of transmit antennas disposed to transmit a set ofspatially-multiplexed RF output signals through a channel using two ormore transmit RF chains and a receiver having a plurality of receiveantennas disposed to receive the set of spatially-multiplexed RF outputsignals and to responsively generate a set of spatially-multiplexedreceived RF signals processed by two or more receive RF chains, anantenna selection method may include, for example, one or more of thefollowing: establishing possible subsets of the plurality of transmitantennas and possible subsets of the plurality of receive antennas;determining one or more output bit error rates of the receivercorresponding to combinations of ones of the possible subsets of theplurality of transmit antennas and ones of the possible subsets of theplurality of receive antennas; identifying a minimum of the one or moreoutput bit error rates; and selecting one of the possible subsets of theplurality of transmit antennas and one of the possible subsets of theplurality of receive antennas collectively corresponding to the minimumof the one or more output bit error rates; and connecting the two ormore of the transmit RF chains to the selected one of the possiblesubsets of the plurality of transmit antennas and the two or morereceive RF chains to the selected one of the possible subsets of theplurality of receive antennas.

In yet another embodiment according to some aspects the presentinvention, in a communication system including a transmitter having aplurality of transmit antennas disposed to transmit a set of RF outputsignals through a channel using one or more transmit RF chains and areceiver having a plurality of receive antennas disposed to receive theset of RF output signals and to responsively generate a set of receivedRF signals processed by one or more receive RF chains, an antennaselection method may include, for example, one or more of the following:establishing possible subsets of the plurality of transmit antennas andpossible subsets of the plurality of receive antennas; determining oneor more output bit error rates of the receiver corresponding tocombinations of ones of the possible subsets of the plurality oftransmit antennas and ones of the possible subsets of the plurality ofreceive antennas; identifying a minimum of the one or more output biterror rates; selecting one of the possible subsets of the plurality oftransmit antennas and one of the possible subsets of the plurality ofreceive antennas collectively corresponding to the minimum of the one ormore output bit error rates; and connecting the one or more of thetransmit RF chains to the selected one of the possible subsets of theplurality of transmit antennas and the one or more receive RF chains tothe selected one of the possible subsets of the plurality of receiveantennas.

In yet another embodiment according to some aspects of the presentinvention, in a receiver having one or more RF chains and a plurality ofreceive antennas capable of receiving transmitted RF signal energy, anantenna selection method may include, for example, one or more of thefollowing: connecting the one or more RF chains to an initial subset ofthe plurality of receive antennas; determining a bit error rateassociated with each receive antennas within the initial subset of theplurality of receive antennas; comparing each the bit error rate to apredefined threshold; disconnecting, when one the bit error rate exceedsthe predefined threshold, an associated receive antenna within theinitial subset of the plurality of receive antennas from a first of theone or more RF chains; and connecting one of the first plurality ofreceive antennas to the first of the one or more RF chains, the one ofthe first plurality of receive antennas not being included within theinitial subset of the plurality of receive antennas.

In yet still another embodiment according to some aspects of the presentinvention, in a transmitter having one or more RF chains and a pluralityof transmit antennas capable of transmitting RF signal energy to areceiver, an antenna selection method may include, for example, one ormore of the following: connecting the one or more RF chains to aninitial subset of the plurality of transmit antennas; determining a biterror rate of the receiver associated with each transmit antennas withinthe initial subset of the plurality of transmit antennas; comparing eachthe bit error rate to a predefined threshold; disconnecting, when onethe bit error rate exceeds the predefined threshold, an associatedtransmit antenna within the initial subset of the plurality of transmitantennas from a first of the one or more RF chains; and connecting oneof the first plurality of transmit antennas to the first of the one ormore RF chains, the one of the first plurality of transmit antennas notbeing included within the initial subset of the plurality of transmitantennas.

BRIEF DESCRIPTION OF THE DRAWINGS

FIGS. 1A-B show an embodiment of a conventional MIMO communicationsystem.

FIGS. 2A-B show an embodiment of a MIMO system according to some aspectsof the present invention.

FIG. 3 shows an embodiment of an SM-MIMOOFDM system according to someaspects of the present invention.

FIGS. 4A-B show flowcharts illustrating embodiments of an antennaselection method according to some aspects of the present invention.

FIG. 5 shows a graph illustrating exemplary performance resultsaccording to some aspects of the present invention.

FIG. 6 shows an embodiment of an SC-MIMO-OFDM system according to someaspects of the present invention.

FIG. 7 shows an embodiment of a receiver in a DS-SS SIMO systemaccording to some aspects of the present invention.

DETAILED DESCRIPTION OF THE INVENTION

I. Overview of Some Aspects of the Present Invention

Some embodiments according to some aspects of the present invention mayrelate to communication systems that use a transmitter and/or a receiverthat have multiple antenna elements.

Some embodiments according to some aspects of the present invention maybe implemented to facilitate the selection of a subset of antennaelements in one or more multi-antenna wireless communication devices soas to minimize or to optimize a bit error rate (BER).

Some embodiments according to some aspects of the present invention maybe employed to select a subset of antenna elements of a multi-antennatransmitter to transmit a signal and/or to select a subset of antennaelements of a multi-antenna receiver to receive a signal.

Some embodiments according to some aspects of the present invention mayprovide that the selection of the subset of antenna elements ispredicated, at least in part, upon a ininmization of a BER.

Some embodiments according to some aspects of the present invention maybe used for antenna selection in a multiple-input-multiple-output (MIMO)communication system. The MIMO communication system may provide, forexample, a transmitter that broadcasts a plurality (N) ofspatially-multiplexed signals through N transmit antenna elementsselected from a set of n_(T) antenna elements, where n_(T)>N. The MIMOcommunication system may provide, for example, a receiver in which Nreceive antenna elements, selected from a total of M elements where M>N,form a number of output signals equal to the number ofspatially-multiplexed signals. The output signals are in turn providedto corresponding RF chains for processing at baseband. Thus, someembodiments according to some aspects of the present invention mayadvantageously permit minimization of BER and/or reduction of RF signalprocessing costs within multiple-antenna systems.

Some embodiments of an antenna selection method according to someaspects of the present invention may be used with different types ofmulti-antenna communication systems. In particular embodiments, someembodiments of an antenna selection method according to some aspects ofthe present invention may be applied, for example, to a multi-antennareceiver within a “single channel” (SC) system (e.g., a system withoutspatial multiplexing), to a multi-antenna transmitter in a singlechannel system, or to the transmitter and/or receiver of a MIMO systememploying spatial multiplexing (SM) or single channel.

Some embodiments according to some aspects of the present invention mayprovide, for example, a number N of receive antenna elements that areselected from a set of M available antenna elements (where M>N) suchthat the selected subset of antenna elements minimizes the BER. This maybe effected, for example, by first estimating the BER of all possiblemulti-antenna subsystems, each of which is comprised of a subset of theM antenna elements. Since the BER is generally a complicated functionof, for example,. the applicable communication channel, in certainimplementations BER may be approximated for a given channel and/orantenna combining technique such that it varies with the coding and/ormodulation method employed. Once the BER is estimated for eachmulti-antenna subsystem, the one yielding the minimum value of BER isidentified. The receive antenna elements of the identified subset arethen connected to multiple RF chains of the receiver to enable receptionof the incoming wireless signal.

A substantially similar approach may be used to select a subset of Ntransmit antenna elements within a multi-antenna transmitter structurecomprising n_(T) transmit elements, where n_(T)>N. Again, the BER may beestimated at the receiver with respect to each of the possiblemulti-antenna subsystems existing at the transmitter. The receiver theninforms the transmitter (e.g., via a feedback path) of the identity ofthe subsystem yielding the minimum BER. This subset is then connected tothe multiple RF chains of the transmitter in order to enabletransmission of the wireless signal. A similar approach may be used forantenna selection when multiple antennas are used at both transmitterand receiver in a single-channel multiple-input-multiple-output (MIMO)system.

Some embodiments according to some aspects of the present invention maybe used for antenna selection in a MIMO communication system having atransmitter operative to broadcast a number (N) of spatially-multiplexedsignals using a subset of N transmit antenna elements selected fromn_(T) available transmit antenna elements, where n_(T)>N. In this casethe receiver uses a subset of N receive antenna elements selected from aset of M receive antenna elements (where M>N). The resulting set of Noutput signals are then fed to a corresponding RF chain for processingat baseband. Some embodiments according to some aspects of the presentinvention thus enable minimization of output BER and RF processing costswithin multi-antenna systems.

In the case of a single-channel or spatially-multiplexed MIMO systemwhich uses multiple RF chains at transmit and/or receiver side, certainbaseband weighting and combining arrangements may be incorporated withinthe transmitter (e.g., precoding) and/or receiver together with theselection method. For example, the baseband weights and antennaselection may be both designed such that they contribute to minimize theBER. In another example, the baseband weights may be designed tomaximize, for example, an output signal-to-noise ratio (SNR), asignal-to-interference-and-noise ratio (SINR), or a capacity whileminimizing BER through appropriate antenna selection.

Some embodiments according to some aspects of the present invention mayalso be employed in systems lacking the capability to determine the BERassociated with all of the antenna elements of a given transmitterand/or receiver in the manner described above. For example, certainsystems may not be equipped with a training sequence structurepermitting training signature information to be collected “in parallel”for each of the antennas of a particular device for the same channelrealization. In this case, the BER of a subset of the total number ofantennas (e.g., the active antennas) is estimated and compared to apre-selected threshold. As soon as the BER of one or more of the activeantennas exceeds the threshold, another antenna is selected andconnected to the RF chain to replace it. This approach may be employedto facilitate antenna selection within transmitters and/or receivers aswell within the context of any of the multi-antenna environmentsdescribed herein.

Some embodiments according to some aspects of the present invention aredirected to a method for antenna selection based upon minimizing the BERfor use in multi-antenna systems, including, for example, N-foldspatially-multiplexed multi-antenna systems. To facilitate appreciationof some embodiments according to some aspects of the present invention,an overview is provided of exemplary architectures for implementingantenna selection within multi-antenna systems.

II. Architecture for Antenna Selection

Some embodiments according to some aspects of the present invention maybe implemented in wireless communication systems in which a smallernumber of RF chains are used within a transmitter and/or receiver thanthe number of transmit/receiver antennas utilized. In some embodimentsaccording to some aspects of the present invention, a number N ofreceive antenna elements is selected out of a total number of elementsM, where M>N. This forms N RF output signals, which are then passedthrough N RF chains. In an exemplary implementation, each RF chainincludes, for example, a filter, downconverter, and A/D converter. Theoutput signals produced by the A/D converter of each RF chain are thendigitally processed to generate the N spatially-multiplexed outputsignals. By performing the requisite selection of a subset of antennasat RF, an N-fold spatially-multiplexed system having more than N receiveantennas, but only N RF chains, can be realized at a cost similar tothat of a system having N receive antennas. Accordingly, receiverperformance may be improved through use of additional antennas atrelatively low cost.

A similar technique can be used at a transmitter incorporating N RFchains and a number n_(T) of transmit antennas that is greater than N.In an exemplary implementation the N RF chains are followed by a switchwhich connects each of them to a subset of N transmit antennas selectedout of n_(T). As at the receiver, by performing such selection of asubset of antennas at RF, an N-fold spatially-multiplexed system havingmore than N transmit antennas, but only N RF chains, can be realized ata cost similar to that of a system having N transmit antennas and N RFchains. Accordingly, transmitter performance may be improved through useof additional antennas at relatively low cost.

A. Spatial Multiplexing

According to some embodiments according to some aspects of the presentinvention, spatial multiplexing (SM) provides a mode of signaltransmission predicated upon the use of multiple antennas at both atransmitter and a receiver in such a way that the bit rate of a wirelessradio link may be increased without correspondingly increasing power orbandwidth consumption. In the case in which N antennas are used at botha transmitter and a receiver, an input stream of information symbolsprovided to the transmitter is divided into N independent substreams.Spatial multiplexing contemplates that each of these substreams willoccupy the same “channel” (e.g., a time slot, a frequency or a code/keysequence) of the applicable multiple-access protocol. Within thetransmitter, each substream is separately applied to the N transmitantennas and propagated over an intervening multipath communicationchannel to a receiver. The composite multipath signals are then receivedby a receive array of N receive antennas deployed at the receiver. Atthe receiver, a “spatial signature” defined by the N phases and Namplitudes arising at the receive antenna array for a given substream isthen estimated. Signal processing techniques are then applied in orderto separate the received signals, which permit the original substreamsto be recovered and synthesized into the original input symbol stream.The principles of spatially-multiplexed communication and exemplarysystem implementations are further described in, for example, “Optimumcombining for indoor radio systems with multiple users”, by J. H.Winters, EEE Transactions on Communications, vol. COM-35, no. 11,November 1987, which is hereby incorporated herein by reference in itsentirety.

B. Conventional MIMO System

Some aspects of the present invention may be more fully elucidated byfirst considering a conventional MIMO communication system, which isillustratively represented by FIG. 1. As shown, the MIMO system 100 ofFIG. 1 includes a transmitter 110 depicted in FIG. 1A and a receiver 130depicted in FIG. 1B. The transmitter 110 and receiver 130 include a setof T transmit RF chains and a set of R receive RF chains, respectively,which are configured to transmit and receive a group of Nspatially-multiplexed signals. Within the system 100 it may be assumedthat either (i) T is greater than N and R is equal to N; (ii) T is equalto N and R is greater than N; or (iii) both T and R are greater than N.

Referring to FIG. 1A, an input signal S to be transmitted, whichtypically includes of a stream of digital symbols, is demultiplexed bydemultiplexer 102 into N independent substreams S_(1, 2, . . . N). Thesubstreams S_(1, 2 . . . , N) are then sent to digital signal processor(DSP) 105, which generates a set of T output signals T_(1, 2 . . . , T).The T output signals T_(1, 2 . . . , T) are typically generated from theN substreams S_(1, 2 . . . , N) by weighting (e.g., multiplying by acomplex number) each of the N substreams S_(1, 2 . . . , N) by Tdifferent weighting coefficients to form NT substreams. These N·Tsubstreams are then combined in order to form the T output signalsT_(1, 2 . . . , T). The T output signals T_(1, 2 . . . , T) are thenconverted to T analog signals A_(1, 2 . . . , T) using a set of Tdigital-to-analog (D/A) converters 108. Each of the T analog signalsA_(1, 2 . . . , T) is then upconverted to the applicable transmitcarrier RF frequency within a mixer 112 by mixing with a signal providedby a local oscillator 114. The resulting set of T RF signals (e.g.,RF_(1, 2 . . . , T)) are then amplified by respective amplifiers 116 andtransmitted by respective antennas 118.

Referring now to FIG. 1B, the RF signals transmitted by the transmitter110 are received by a set of R receive antennas 131 deployed at thereceiver 130. Each of the R signals received by an antenna 131 isamplified by a respective low noise amplifier 133 and passed through afilter 135. The resultant filtered signals are then each downconvertedfrom RF to baseband using mixers 137, each of which is provided with asignal from local oscillator 138. Although the receiver of FIG. 1B isconfigured as a homodyne receiver, a heterodyne receiver characterizedby an intermediate IF frequency could also be used. The respective Rbaseband signals produced by the mixers 137 are then converted todigital signals using a corresponding set of R analog-to-digital (A/D)converters 140. The resulting R digital signals D_(1, 2. . . . , R) arethen weighted and combined using digital signal processor 142 to form Nspatially-multiplexed output signals S′_(1, 2 . . . , N) which compriseestimates of the transmitted signals S_(1, 2 . . . , N). The N outputsignals S′_(1, 2 . . . , N) are then multiplexed using a multiplexer 155in order to generate an estimate 160 (S′) of the original input signalS.

C. Antenna Selection at RF in Spatially-Multiplexed CommunicationSystems

Turning now to FIG. 2, there is shown a block diagram of a MIMOcommunication system 200 having a transmitter 210 and receiver 250configured to effect N-fold spatial multiplexing using only Ntransmit/receive RF chains, even though more than N transmit/receiveantennas are respectively deployed at the transmitter 210 and receiver250. Specifically, the transmitter 210 includes a set of MT transmitantennas 240 and the receiver includes a set of MR receive antennas 260,some embodiments according to some aspects of the present invention mayprovide that MT and/or MR are greater than or equal to N. For example,(i) MT is greater than N and MR is equal to N; (ii) MT is equal to N andMR is greater than N; or (iii) both MT and MR are greater than N.

As shown in FIG. 2A, an input signal S to be transmitted isdemultiplexed by demultiplexer 202 into N independent substreamsSS_(1, 2 . . . , N). The substreams SS_(1, 2 . . . , N) are thenconverted to N analog substreams AS_(1, 2 . . . , N) using acorresponding set of D/A converters 206. Next, the N analog substreamsAS_(1, 2 . . . , N) are upconverted to the applicable transmit carrierRF frequency using a set of mixers 212 provided with the signal producedby a local oscillator 214. The resultant N RF signals (e.g.,RF_(1, 2 . . . , N)) are then each connected to a selected subset of Ntransmit antenna elements by a switch 218. The switch 218 connects N RFsignals (e.g., RF_(1, 2 . . . , N)) to a set of N transmit antennas fromthe MT available transmit antennas 240, thereby yielding a set of Noutput signals. A corresponding set of N amplifiers 234 then amplifythese N output signals, with the amplified output signals then beingtransmitted using the N selected transmit antennas 240. In anotherexample, the amplifiers 234 may be located before the switch 218. Inthis configuration, a total of only N amplifiers is needed instead of atotal of MT if one amplifier is placed at each of the MT antennas. Theselection of the N antennas is generated so as to minimize the BER ofthe output signal at the receiver.

Referring to FIG. 2B, the N RF signals transmitted by the transmitter210 are received by the set of MR receive antennas 260 deployed at thereceiver 250. Each of the MR received signals is amplified by arespective low noise amplifier (LNA) 264 and then a subset N of them isconnected to a set of N RF chains by a switch 276 in order to form a setof N RF signals which are passed through a corresponding set of Nfilters 280. In another example, the low noise amplifier 264 may belocated after the switch 276 such that the total number of used LNA is Ninstead of MR if one LNA is placed at all MR receive antenna elements.The resulting N filtered signals are then downconverted to basebandusing a set of N mixers 282, each of which is provided with a carriersignal produced by a local oscillator 284. Although the receiver 250 isrealized as a homodyne receiver in the embodiment of FIG. 2B, it couldalso be implemented as a heterodyne receiver characterized by anintermediate IF frequency. (In fact, any of the embodiments according tosome aspects of the present invention may incorporate, for example,homodyne configurations or heterodyne configurations). The N basebandsignals produced by the mixers 282 are then converted to digital signalsvia a corresponding set of N A/D converters 286. The N digital signalsare then feather processed using digital signal processor 288 to formthe N spatially-multiplexed output signals SS′_(1, 2 . . . , N), whichare the estimates of the N independent substreams SS_(1, 2 . . . , N).The N output signals SS′_(1, 2 . . . , N) are then multiplexed via amultiplexer 292 in order to generate the output signal S′, which is anestimate of the input signal S.

In some embodiments according to some aspects of the present invention,a baseband weighting and combining (e.g., a “preceding”) arrangement maybe added at the transmitter side for use in conjunction with the antennaselection method discussed below. In this case a DSP block is placedbetween the demultiplexer 202 and the D/A converters 206, such that theN independent substreams SS_(1, 2 . . . , N) are weighted by complexcoefficients and combined to form a set of N output signals. These Noutput signals are then converted into analog signalsAS_(1, 2 . . . , N) using the corresponding set of D/A converters 206.

In some embodiments according to some aspects of the present invention,space-time coding can be added at the transmitter side for use inconjunction with an exemplary antenna selection method. In this case,the demultiplexer 202 is replaced by a DSP block which processes theinput signal S over the space and time domain to form a set of N outputsignals. These N output signals are then converted into analog signalsAS_(1, 2 . . . , N) using the corresponding set of D/A converters 206.Among the two most commonly used space-time techniques are 1) theinsertion of a time delay (or equivalently a phase-shift) on one or moreof the N output signals and 2) the use of the transmit diversitytechnique described in, for example, “A simple transmit diversitytechnique for wireless communications”, by S. M. Alamouti, IEEE Journalon Selected Areas in Communications, vol. 16, issue 8, October 1998,pages 1451 -1458, which is hereby incorporated herein by reference inits entirety.

Space-time coding techniques may be applicable, for example, to the SCMIMO systems and/or systems designed to yield diversity gain. Precodingtechniques may be applicable, for example, to SC-based orspatial-multiplexing-based MIMO systems or systems designed to yieldboth data rate and diversity gains.

III. Antenna Selection Method at RF Based on Minimum Bit Error Rate

A. Overview

Some embodiments according to some aspects of the present inventionrelate, for example, to an antenna selection method in a multi-antennacommunication system predicated upon minimizing a bit error rate. Insome embodiments according to some aspects of the present invention, asubset of antenna elements is selected to transmit and/or receive thesignals such that the bit error rate is minimized, for example, in acommunication system with multiple antennas. Some embodiments accordingto some aspects of the present invention may be used for antennaselection at the transmitter when multiple antennas are used fortransmission. Some embodiments according to some aspects of the presentinvention can be used for antenna selection at the receiver whenmultiple antennas are used for reception.

Some embodiments according to some aspects of the present invention maybe applicable to, for example, (i) receivers using multiple antennas inwhat are referred to herein as single channel systems (e.g., systemlacking spatial multiplexing); (ii) transmitters using multiple antennasin single channel systems; and (iii) systems in which a smaller numberof RF chains are used at the transmitter and/or receiver than the numberof transmit and/or receiver antennas in a MIMO system with spatialmultiplexing or single-channel.

Some embodiments according to some aspects of the present invention willbe described hereinafter with reference to FIGS. 3-7 within thefollowing exemplary contexts: 1) a MIMO system with spatial multiplexingin which a smaller number of RF chains are used at the transmitter andreceiver than the number of transmitter/receiver antennas; 2) asingle-channel MIMO system without spatial multiplexing in which asmaller number of RF chains are used at the transmitter and receiverthan the number of transmitter/receiver antennas; and 3) asingle-channel SIMO system without spatial multiplexing containing areceiver using multiple antenna elements. Some embodiments according tosome aspects of the present invention may also be employed in the caseof a single-channel (SC) multiple-input-single-output (MISO) systemwithout spatial multiplexing in which a transmitter utilizes multipleantenna elements.

For illustrative purposes, the following exemplary examples aredescribed with reference to systems utilizing OFDM modulation (e.g.,following the 802.1-1a WLAN standard) or to systems based upon a directsequence spread spectrum (DS-SS) (e.g., following the WCDMA standard).In certain embodiments according to some aspects of the presentinvention, the processing capabilities of the DS-SS receiver may beextended to cover the spatial domain through incorporation of aspace-time Rake receiver operative to combine multi-path “taps”corresponding to both the temporal and spatial domains. This extensionillustrates that the techniques described herein may be generalized tovirtually any system employing, for example, temporal and/or frequencydomain processing in a frequency-selective fading environment.

B. Antenna Selection in a SM-MIMO-OFDM System

FIG. 3 illustratively represents the transmitter and receiver structureof an SM-MIMO-OFDM system 300 utilizing antenna selection in accordancewith an embodiment according to some aspects of the present invention.As shown, two independent sub-streams 304 (e.g., spatially-multiplexedsignals) are OFDM-modulated onto N_(t) frequency sub-carriers and passedthrough two RF chains 308 to prepare for transmission. At this point, aswitching block 312 selects two of four transmit antenna elements 316 toconnect to the two RF chains 308. Since only two out of four elements316 are selected within the transmitter 302, the number of transmit RFchains is advantageously reduced to the number of spatially-multiplexedsignals.

In the embodiment of FIG. 3, the switching block 312 containsinformation identifying the pair of antenna elements 316 to be used fortransmission at any given time. The block 312 may compute thisinformation itself (e.g., in the case where the channel 318 isreciprocal) in accordance with an algorithm based upon the minimum BERcriterion. In another example, the block 312 may receive the informationfrom the receiver 330 via a feedback path (not shown). This latterapproach may be used in the case where the channel 318 is notreciprocal, for example, in an interference-limited environment.

Within the receiver 330, a switching block 334 selects two of fourantenna elements 338 to receive incident signals transmitted by thetransmitter 302. The switching block 334 connects the two selectedantennas 338 to two RF chains 342 operative to convert the two signalsinto the digital domain for baseband processing. Then, a weight matrix346 is applied to the received signals at each tone to separate andrecover each one of the transmitted spatially-multiplexed signals.

In typical implementations the switching block 334 will be configured toitself compute which pair of antenna elements 338 should be selected forreception by executing an algorithm based upon the minimum BERcriterion. In the case where the channel is not reciprocal, the block334 may be further configured to compute which pair of antenna elements316 should be used in the transmitter 302 and to provide thisinformation to it. A description of two possible implementations of anantenna selection algorithm executable by the switching blocks 312, 334is provided with reference to FIGS. 4A and 4B.

Turning now to FIG. 4A, a flowchart is provided of an antenna selectionalgorithm 400 in which the coding/modulation mode (e.g., data rate orthroughput) is fixed or adapted on a long-term basis (e.g., adapted tothe large-scale variations of the SNR). The task of the selectionalgorithm is to select which pair of antenna elements 316 should be usedat the transmitter 302 and which pair of antenna elements 338 should beused at the receiver 330 with respect to each packet for the given mode.The selection process may assume, for example, that the channel 318 isquasi-stationary (e.g., the channel 318 is constant over the duration ofthe packet being transmitted and changes independently between twocontiguous packets). Even though the channel 318 may exhibit somefrequency selectivity, the antenna selection may be common to the entirefrequency bandwidth.

Referring to FIG. 4A, when the transmitter 302 initially powers up (step401) and the state of the channel 318 is still unknown, a default subsetof two of the antenna elements 316 is used to transmit the wirelesssignal. The receiver 330 similarly uses a default subset of two of thereceive antenna elements 338 in order to acquire synchronization. Next,channel state information (CSI) is acquired (step 402). In someembodiments according to some aspects of the present invention,operations to acquire CSI are carried out at the receiver 330. Atraining sequence composed of known symbols is sent from the transmitter302 to the receiver 330. At the receiver 330, the channel 318 isestimated based on the received signal and the known sequence ofsymbols. This operation is carried out as often as the channel 318changes, for example, at each packet realization. In order for theselection method to be performed successfully, the complete channelmatrix should be estimated over the whole frequency bandwidth (e.g., theestimation of the channel path gain from all antenna elements 316 of thetransmitter 302 to all antenna elements 338 of the receiver across alltones). Channel estimation techniques based on training sequencesapplicable to MIMO systems are described in, for example, J. J. Van deBeek et al., “On Channel Estimation in OFDM Systems”, IEEE 45thVehicular Technology Conference, vol. 2 , 25-28 Jul. 1995, pp. 815-819and A. N. Mody and G. L. Stuber, “Synchronization for MIMO OFDMSystems”, IEEE Globecom 2001, vol. 1, pp. 509-513, which are herebyincorporated herein by reference in their entirety.

Referring again to FIG. 4, mode information is acquired throughexecution of a link adaptation algorithm (step 404). In the embodimentillustrated by FIG. 4A, the mode change may occur slowly. This enables alink adaptation algorithm to be employed to decide which of the possiblemode candidates is best suited to be used in view of the long-termaverage SNR. Employment of a link adaptation algorithm may ensure that,given a mode selection criterion (e.g., a maximum data rate and aminimum transmit power), the most efficient mode is always used in viewof long-term varying channel/SNR conditions. An exemplary linkadaptation algorithm capable of being utilized withinfrequency-selective MIMO systems is described, for example, in “AdaptiveModulation and MIMO Coding for Broadband Wireless Data Networks”, by S.Catreux et al., IEEE Communications Magazine, vol. 40, No. 6, June 2002,pp. 108-115, which is incorporated herein by reference in its entirety.The mode selection may generally be independent of the method ofselecting transmitter/receiver antenna elements. The mode may beselected based exclusively upon long-term SNR statistics. Accordingly,it changes at a much slower rate than that at which the antennas areselected. In other words, the selection algorithm may select a newsubset of antennas with respect to each packet. realization, while themode changes as a function of long-term SNR variations.

Steps 406, 408 and 410 are repeatedly executed in a loop until allpossible combinations of subsets of transmit/receive antenna elementshave been evaluated (step 411). For example, considering a MIMO-OFDMsystem of the type depicted in FIG. 3 (e.g., equipped with 4 transmitantenna elements 316 and 4 receive antenna elements 338), the completechannel matrix can be represented in the frequency domain at tone k by a4×4 matrix denoted by H_(k). After selection of a subset of two antennasat each side, the sub-channel matrix is reduced in size to a 2×2 matrixdenoted by {tilde over (H)}_(k). There are

$\begin{pmatrix}4 \\2\end{pmatrix} = 6$possibilities in selecting 2 elements out of a total of 4. Since theantenna selection is applied at both the transmitter 302 and receiver330, the total number of combinations possible for {tilde over (H)}_(k)is equal to 36. In the general case of a M×M MIMO system being reducedin size to a n×n MIMO system (where M>n), there are

$\begin{pmatrix}M \\n\end{pmatrix} = \frac{M!}{{n!}{( {M - n} )!}}$possibilities in selecting n antenna element from M possible elements.When the selection occurs at both a transmitter and receiver, the totalnumber of combinations for {tilde over (H)}_(k) is equal to

$( \frac{M!}{{n!}{( {M - n} )!}} )^{2}.$This corresponds to the number of iterations of the loop comprised ofsteps 406, 408 and 410. These iterations may be performed in series(e.g., reusing, common processing resources) or in parallel (e.g., atthe expense of additional processing resources). In an exemplaryexample, all possible antenna combinations could be contemporaneouslyprocessed, which might employ a separate processing resource for eachpossible antenna combination.

Each iteration in the loop comprised of steps 406, 408 and 410 affectsprocessing of one antenna subsystem. First, the 2×2 matrix {tilde over(H)}_(k) is acquired across all tones (k=1, . . . , N₁) for thesubsystem of interest (Step 406). The post-processingsignal-to-interference-and-noise ratio (SINR) is then computed at eachtone k and for each transmitted spatially-multiplexed signal (Step 408).The SINR can most often be found by a closed-form solution dependentupon which signal processing technique is used at the transmitter 302and/or receiver 330 (e.g., Maximum Ratio Combining (MRC), Minimum MeanSquare Error (MMSE), eigen-beamforming, and Maximum Likelihood (ML)).For example, if no spatial processing is implemented at the transmitter302 and MMSE is applied at the receiver 330, the SINR may be determinedas follows:

-   -   Compute

$B_{k} = {{{\overset{\sim}{H}}_{k}^{H}{\overset{\sim}{H}}_{k}} + {\frac{\sigma^{2}}{\sigma_{s}^{2}}I_{2}}}$with ${I_{2} = \begin{bmatrix}1 & 0 \\0 & 1\end{bmatrix}},\sigma^{2}$and σ_(s) ² stand for noise and signal power respectively and k=1, . . ., N_(t) (step 408-1).

-   -   Compute C_(k)=1/diag(B_(k) ⁻¹) this is a N×1 vector for each        k=1, . . . , N_(t) (step 408-2).    -   Compute

${SINR}_{K} = {{\frac{\sigma_{s}^{2}}{\sigma^{2}}C_{k}} - 1}$this is a N×1 vector for each k=1, . . . , N_(t) (step 408-3).

In step 410, the SINR information is converted into BER information inview of the current mode (see, e.g., step 404). Since the BER may be acomplicated function of the channel 318 and of the coding/modulation andantenna combining techniques used, an approximation of the BER may beused. The approximation may also be a function of the channel 318 and ofthe applicable coding/modulation and antenna combining techniques. TheBER over the packet (e.g., at the output of the Viterbi decoder ifcoding is used) for transmitted substream i may be expressed as anon-linear, unknown function ƒ of the set of SINR_(k), k=1, . . . ,N_(t), for example:BER _(i)=ƒ({SINR_(k) ^(i)}), i=1, . . . , N; k=1, . . . , N_(t)

Next, the function ƒ is approximated by some known function.Specifically, the output BER is approximated by the average of the biterror rate over the channel, for example:

$\begin{matrix}{{\overset{\_}{BER}}_{i} \approx {{1/N_{t}}{\sum\limits_{k = 1}^{N_{t}}{{BER}_{k}^{i}.}}}} & (1)\end{matrix}$where BER_(k) ^(i) is the bit error rate given the SINR at tone k forspatially-multiplexed substream i. In another example, BER_(k) can alsobe the bit error rate given the signal-to-noise ratio at tone k. Theaverage may also be taken in the time domain where BER_(k) is the biterror rate given the SINR at channel time sample k. BER_(k) may be thebit error rate with respect to a given signal component (e.g., a signaltone or tap delay).

Furthermore, BER_(k) is also approximated by some simple closed-formfunction. Through simulations it has been found that for mode 1 of802.1-1a (e.g., BPSK, R1/2), the behavior of the average bit error rateBER with respect to the SINR or SNR (in some examples, the BERnormalization factor 1/N_(t) may be omitted, since it does not affectthe antenna selection) can be modeled, for example, by:

$\begin{matrix}{{{{\overset{\_}{BER}}_{i} \approx {- {\sum\limits_{k = 1}^{N_{t}}{\tan\;{h( {SINR}_{k}^{i} )}}}}};{i = 1}},\ldots\mspace{14mu},{N.}} & (2)\end{matrix}$The BER at signal component k has been approximated by −tanh (SINR_(k)).

The tanh function may not always adequately approximate the BER,particularly for different modulation techniques. Some of the followingfunctions may generally afford better approximations when usingparticular techniques:

1) The BER of an uncoded BPSK modulation in AWGN channel is (see, e.g.,J. G. Proakis, Digital Communications, 3^(rd) Ed. McGraw-Hill Series,1995)

${BER}_{BPSK} = {{Q( \sqrt{\frac{2E_{b}}{N_{o}}} )} = {{Q( \sqrt{2\gamma_{b}} )} = {{\frac{1}{2}{{erfc}( \sqrt{\gamma_{b}} )}} = {\frac{1}{2}{{{erfc}( \sqrt{\gamma_{s}} )}.}}}}}$The shape of the function erfc is reasonably approximated by thefunction (as compared to y=−tanh(x))y=−[(1−e ^(−2√{square root over (x)}))+(1−e ^(−1.8x))]

2) The BER of an uncoded QPSK modulation in AWGN channel is (see, e.g.,J. G. Proakis, Digital Communications, 3^(rd) Ed. McGraw-Hill Series,1995)

${BER}_{QPSK} = {{Q( \sqrt{\frac{2\; E_{b}}{N_{o}}} )} = {{Q( \sqrt{2\;\gamma_{b}} )} = {{\frac{1}{2}{{erfc}( \sqrt{\gamma_{b}} )}} = {\frac{1}{2}{{{erfc}( \sqrt{\frac{\gamma_{s}}{2}} )}.}}}}}$The shape of erηc(√{square root over (x/2)}) is better approximated bythe function:y=−[(1−e ^(−1.3√{square root over (x)}))+(1−e ^(−x))]than by y=−tanh(x).

3) The BER of an uncoded 16QAM modulation in AWGN channel may be derivedfrom the symbol error rate (SER) given, for example, in J. G. Proakis,Digital Communications, 3^(rd) Ed. McGraw-Hill Series, 1995 as

${BER}_{16{QAM}} = {{1 - \sqrt{1 - {\frac{3}{2}{Q( \sqrt{\frac{3\; E_{s}}{15\; N_{o}}} )}}}} = {1 - \sqrt{1 - {\frac{3}{4}{{erfc}( \frac{\gamma_{s}}{10} )}}}}}$An appropriate fitting function is y=−(1−e ^(−0.2x)).

4) The BER of an uncoded 64QAM modulation in AWGN channel may be derivedfrom the symbol error rate (SER) given, for example, in J. G. Proakis,Digital Communications, 3^(rd) Ed. McGraw-Hill Series, 1995 as

${BER}_{64{QAM}} = {{1 - ( {1 - {\frac{7}{4}{Q( \sqrt{\frac{\gamma_{s}}{21}} )}}} )^{1/3}} = {1 - ( {1 - {\frac{7}{8}{{erfc}( \sqrt{\frac{\gamma_{s}}{42}} )}}} )^{1/3}}}$An appropriate fitting function isy=−(1−e^(−0.35√{square root over (x)})).

It is to be understood than any fitting function that reasonably modelsthe behavior of the BER versus SINR can be used in equation (2). Thenumber of appropriate fitting functions is not limited to the fewexamples given above.

As mentioned above, steps 406 to 410 are iteratively performed until allpossible combinations of subsets of antennas are considered (step 411).At the conclusion of this iterative process, a set of N estimates of BERvalues (one for each spatially-multiplexed signal) for all

$\frac{M!}{{n!}{( {M - n} )!}}( {{or}\mspace{11mu}( \frac{M!}{{n!}{( {M - n} )!}} )^{2}} )$possible antennas combinations is obtained. It then remains to selectthe subset of antennas that minimize the mean over the set of BERs, themaximum over the set of BERs or, the minimum of the set of BERs (step412).

$\min\limits_{\underset{combinations}{antennas}}\{ {\underset{{i = 1},\ldots\mspace{14mu},N}{mean}\{ {BER}_{i} \}} \}$$\min\limits_{\underset{combinations}{antennas}}\{ {\min\limits_{{i = 1},\ldots\mspace{14mu},N}\{ {BER}_{i} \}} \}$$\min\limits_{\underset{combinations}{antennas}}\{ {\max\limits_{{i = 1},\ldots\mspace{14mu},N}\{ {BER}_{i} \}} \}$

FIG. 4B is a flowchart of an antenna selection algorithm 500 in whichthe coding/modulation mode may be changed as frequently as once perpacket realization in response to corresponding changes in the channel318. In this example, the coding/modulation mode is adapted at the samerate as antenna selection is effected.

Referring to FIG. 4B, steps 501 and 502 are similar to steps 401 and402, respectively. As shown, steps 504 through 510 comprise a loop thatis iteratively executed until all possible combinations of subsets ofantennas have been evaluated. It follows that the number of iterationsof this loop is equivalent to

$\frac{M!}{{n!}{( {M - n} )!}}$(e.g., selection at one end of the link) or

$( \frac{M!}{{n!}{( {M - n} )!}} )^{2}$(e.g., selection at both ends of the link). In this regard steps 504 and506 are similar to steps 406 and 408, respectively. Based upon theknowledge of the instantaneous SINR at all tones, a link adaptationblock determines the most efficient mode for each spatially-multiplexedsignal, given a mode selection criterion (e.g., a maximum data rate anda minimum transmit power) (step 508). This step is similar to step 404,with the exception that the mode decision is made based uponinstantaneous SNR (or SINR) statistics rather than upon long-term SNR(or SINR) statistics. As a result, different combinations of subsets ofantennas may yield different mode decisions. Finally, given theinstantaneous SINR and mode information, step 510 computes or determinesthe corresponding BER in the same manner as was described above withreference to step 410.

Again, steps 504 through 510 are performed until all possiblecombinations of subsets of antennas are considered (step 511). Once thishas occurred, a set of N estimates of BER values (e.g., one for eachspatially-multiplexed signal) for all

$\frac{M!}{{n!}{( {M - n} )!}}( {{or}\mspace{11mu}( \frac{M!}{{n!}{( {M - n} )!}} )^{2}} )$possible antennas combinations is obtained. The selection algorithm 500differs from the algorithm 400 in that the

$\frac{M!}{{n!}{( {M - n} )!}}( {{or}\mspace{11mu}( \frac{M!}{{n!}{( {M - n} )!}} )^{2}} )$possible antennas combinations do not necessarily use the samecoding/modulation mode. The decision of which antennas subset to selecttherefore depends not only on a minimization of the BER but also on themode (e.g., data rate or throughput). Several exemplary options areprovided with regard to the final decision of selecting a subset ofantenna elements pursuant to step 512 of the selection algorithm 500:

Option 1

-   -   1) Group all combinations of subsets of antennas using the same        mode into a common pool.    -   2) Choose the pool corresponding to the highest mode (yielding        maximum data rate).    -   3) Within that pool, select the combination of subset of        antennas that minimizes the BER in a manner substantially        similar to that described with reference to Step 412.

Option 2

-   -   Regardless of which mode is used by each combination, select the        combination of antenna subsets that minimizes the BER in a        manner substantially similar to that described with reference to        Step 412.

Option 3

-   -   Implement a hybrid version of option 1 and 2, for example:    -   1) Group all combinations of subsets of antennas using the same        mode into a common pool.    -   2) Choose the X pools corresponding to the X highest modes        (yielding maximum data rate), where X is an integer equal to 1,        or 2 or 3, etc.    -   3) Within these pools, select the combination of subset of        antennas that minimizes the BER in a manner substantially        similar to that described with reference to step 412.

FIG. 5 illustratively represents the packet error rate (PER) as functionof SNR resulting from employment of an exemplary antenna selectiontechnique within a SM-MIMO-OFDM system operative in a noise-limitedenvironment. The results of FIG. 5 may be applicable, for example, to asystem using four transmit and receive antennas in the mannerexemplified by FIG. 3. The results reflect, as merely exemplaryexamples, a packet size of 1000 bytes and a fixed coding/modulationmode. The results also reflect that two exemplary RF chains areincorporated within both the applicable transmitter and receiver. Inaddition, the results of FIG. 5 use BPSK modulation, a coding rate of ½(e.g., mode 1 of 802.11a), a channel model characterized as “channel A”(e.g., 50 ns rms delay spread, 0.5 antenna correlation), and a fittingfunction of tanh.

The legend for the curves in FIG. 5 is as follows:

2×2 2SM-MIMO MMSE: This system corresponds to a SM-MIMO-OFDM systemusing 2 transmit and 2 receive antennas together with 2spatially-multiplexed (SM) signals. Since the number of antennas isequal to the number of SM signals, no antenna selection is applied. Abaseband combining arrangement is used at the receiver to separate thetwo substreams, e.g., MMSE.

4×4 2SM-MIMO sel mcap MMSE: This system corresponds to a SM-MIMO-OFDMsystem using 4 transmit and 4 receive antenna elements together with 2spatially-multiplexed (SM) signals. A conventional selection method isapplied at both the transmitter and receiver to select a subset of 2antenna elements among four, according to a maximum capacity criterion.After the selection at the receiver, MMSE is applied at baseband toseparate the two substreams.

2×4 2SM-MIMO sel mber MMSE (bound): This system corresponds to aSM-MIMO-OFDM system using 2 transmit and 4 receive antenna elementstogether with 2 spatially-multiplexed (SM) signals. A selection methodis applied only at the receiver end to select a subset of 2 antennaelements among four, according to the minimum BER criterion. In thiscase, no fitting function is used to approximate the BER. Instead, theBER is assumed to be known perfectly. This case may not be readilyimplemented, but rather provides a bound on the performance which may beachieved through use of some embodiments according to some aspects ofthe present invention.

4×4 2SM-MIMO sel mber MMSE (bound) This system corresponds to aSM-MIMO-OFDM system using 4 transmit and 4 receive antenna elementstogether with 2 spatially-multiplexed (SM) signals. A selection methodis applied at both transmit and receive sides to select a subset of 2antenna elements among four, according to the minimum BER criterion. Inthis case, no fitting function is used to approximate the BER. Instead,the BER is assumed to be known perfectly. This case may not be readilyimplemented, but rather provides a bound on the performance which may beachieved through use of some embodiments according to some aspects ofthe present invention.

4×4 2SM-MIMO sel mber MMSE (implementation tanh) This system correspondsto a SMMIMO-OFDM system using 4 transmit and 4 receive antenna elementstogether with 2 spatially-multiplexed (SM) signals. A selection methodaccording to some embodiments according of the present invention isapplied at both transmit and receive sides to select a subset of 2antenna elements among four, according to a minimum BER criterion. Thefitting function to approximate the BER is tanh.

The results illustratively represented by FIG. 5 show that all systemsusing some type of antenna selection provide gains relative to systemswith no selection, and that antenna selection based upon minimum BERprovides significantly more gain than selection based upon the maximumcapacity criterion. Specifically, at a PER level of 10e-2 and withantenna selection in accordance with the present invention being appliedat both the transmitter and receiver, a 7.6 dB gain is achieved relativeto a system with no selection and 4.2 dB gain is demonstrated relativeto a system in which selection is based on maximum capacity. Whenselection is applied consistent with some embodiments according to someaspects of the present invention at only the receiver, the resultingperformance is seen to be between that achieved when no selection isemployed and that which occurs when selection is employed at both thetransmitter and receiver. Finally, the performance of the system inaccordance with some embodiments according to some aspects of thepresent invention very closely approaches a theoretical performancebound illustrated by FIG. 5.

C. Antenna Selection in a SC-MIMO-OFDM System

FIG. 6 illustrates a SC-MIMO-OFDM system 600 which utilizes precodingtechniques in addition to an exemplary antenna selection methodaccording to some embodiments of the present invention. In theembodiment of FIG. 6, preceding refers to various baseband weighting andcombining arrangements performed at a transmitter 602. Referring to FIG.6, a single-stream of symbols 604 is weighted by a set of complexcoefficients 608 and combined to produce a set of N output signals whereN refers to the number of RF chains 612 used within the transmitter 602.These N output signals are then passed through the N RF chains 612 inorder to produce N RF signals. These N RF signals are then coupled to acorresponding group of N of M transmit antenna elements 616 via a switch620 and transmitted through a channel 624.

At a receiver 622, a set of N of M receive antenna elements 626 isselected via a switch 630 to receive the incoming signals communicatedthrough a channel 624. The N RF receive signals are then processed by NRF chains 634 and converted to the digital domain for basebandprocessing to recover the original transmitted signal.

Some embodiments according to some aspects of the present invention mayprovide, for example, that the baseband weights 608 and antennaselection method are designed to collectively contribute to theminimization of the BER. Some embodiments according to some aspects ofthe present invention may provide, for example, that the basebandweights 608 may be chosen to maximize the output SNR (or SINR) orcapacity, while the antenna selection is conducted so as to minimize theBER. The right and left singular vectors of the sub-channel matrix{tilde over (H)}_(k) corresponding to the largest singular value may beused to select optimal subsets of transmit antenna elements 616 andreceive antenna elements 626 as well as the appropriate transmitbaseband weights 608 and receive baseband weights 640. Determination ofbaseband weighting values in the context of MIMO systems lacking antennaselection has been described, for example, in J. B. Andersen, IEEEAntennas and Propagation Magazine, vol. 42, no. 2, April 2000, pp.12-16, which is incorporated herein by reference in its entirety.

The exemplary embodiment illustrated in FIG. 6 may be modified byreplacing the baseband weights 608 within the transmitter 602 by aspace-time coding block. In this case, an antenna selection method maybe employed to select the subset of antennas in both the transmitter andreceiver in accordance with some embodiments of the present invention.In addition, the space-time coding block processes the input stream ofsymbols as described in, for example, “A simple transmit diversitytechnique for wireless communications”, by S. M. Alamouti, IEEE Journalon Selected Areas in Communications, vol. 16, issue 8, October 1998,pages 1451 -1458, which is incorporated herein by reference in itsentirety.

D. Antenna Selection in a DS-SS-SIMO System

FIG. 7 depicts a receiver 700 of a DS-SS SIMO system having two receiveantenna elements 704 (n_(R)=2). The receiver 700 integrates RAKEreceiver functionality together with an exemplary antenna selectionprocess. As shown, the receiver 700 is equipped with only a single RFchain 708 disposed to be connected to only one of the two receiveantenna elements 704 at any given time via a switch 712. The choice ofwhich of the two elements 704 to connect to the RF chain 708 is madebased upon the minimum BER criterion. Some embodiments according to someaspects of the present invention may provide that the received signalBER corresponding to each receive antenna element 704 is computed andthe element 704 yielding the minimum value of BER is selected. Since theBER may typically comprise a complicated function of the applicablechannel and of the coding/modulation and antenna combining techniquesused, the BER for a given channel and antenna combining technique isapproximated such that it varies as a function of the coding/modulationmethod used.

Once the optimal one of the antenna elements 704 has been selected, theRAKE receiver behaves in the same way as if it were implemented in asingle-input single-output (SISO) system (e.g.,, one antenna at each endof the link). The RAKE receiver uses a plurality of J correlators 720(e.g., J=2 in FIG. 7), each of which corresponds to one of the first Jseparable multipath components. Each such multipath component isassociated with a time delay τ_(j), j =1, . . . , J, respectively. Theoutput of each correlator 720 (e.g., a finger) is then weighted 730 andcombined 740 to form a single output received signal 750 comprising anestimate of the transmitted signal.

In one exemplary example, the received signal corresponding to thei^(th) antenna element 704 at the input of the RAKE receiver may beexpressed as:

$\begin{matrix}{{r_{i}(t)} = {{\sum\limits_{l = 1}^{L_{i}}{h_{i,l}\sqrt{2P}{d( {t - \tau_{i,l}} )}{p( {t - \tau_{i,l}} )}{\cos( {{w_{0}( {t - \tau_{i,l}} )} - \theta_{i,l}} )}}} + {{n_{i}(t)}.}}} & (3)\end{matrix}$where L_(i) is the number of taps in the channel received at the i^(th)antenna element 704, h_(i,j) is the complex channel gain at antenna iand tap l, P is the signal transmit power, d is the data sequencecomprised of symbols of period T, and p is the spreading sequencecomposed of chips of period T_(c)=T/G, where G is the spreading factor.In addition, τ_(i,l) is the path delay associated with tap l and antennai, w₀ corresponds to the carrier frequency w₀=2πƒ₀, and θ_(i,t) is thephase shift corresponding to tap l and antenna i. The noise n_(i)measured at the i^(th) antenna element 704 is modeled as an AWGN processwith two-side spectral density N₀/2. For the sake of simplicity andclarity of expression, equation (3) assumes a single-user environment.However, the present invention need not be so limited and alsocontemplates being applied in the presence of multiple users.

At the output of the correlator 720 of the j^(th) finger, the receivedsignal may be represented as:

$\begin{matrix}{\begin{matrix}{r_{i,j} = {\sqrt{\frac{2}{T}}{\int_{\tau_{j}}^{\tau_{j}}{{r_{i}(t)}{p( {t - \tau_{j}} )}{\cos\ ( {{w_{0}( {t - \tau_{j}} )} - \theta_{j}} )}{\mathbb{d}t}}}}} \\{= {{\sqrt{PT}h_{i,j}d_{0}} + n_{i,j}}}\end{matrix}.} & (4)\end{matrix}$where d₀ is the desired symbol to be demodulated, and n_(ij) is the AWGNnoise component with zero-mean and with two-side spectral density N₀/2.Again for purposes of simplicity and clarity of presentation, it wasassumed in equation (4) that there is no interpath interference (IPI).However, the present invention also contemplates being used in thepresence of IPI.

Following diversity combining, the final output of the RAKE receivercorresponding to the i^(th) antenna element 704 is:

$\begin{matrix}{r_{i} = {\sum\limits_{j = 1}^{J}{w_{i,j}{r_{i,j}.}}}} & (5)\end{matrix}$where J is the number of RAKE fingers and where the optimum combiningweights are generally chosen so as to match the channel, for example:w_(i,j)=h_(i,j)*   (6.)

In this case, the RAKE performs maximum ratio combining and the SNR atthe RAKE output, corresponding the i^(th) antenna element 704 is givenby

$\begin{matrix}{\gamma_{i} = {\sum\limits_{j = 1}^{J}{\gamma_{i,j}.}}} & (7)\end{matrix}$where γ_(i,j) is the post-combining SNR on the j^(th) path associatedwith the i^(th) antenna element 704. Based on (4), γ_(i,j) may beexpressed by:

$\begin{matrix}{{\gamma_{i,j} = \frac{{h_{i,j}}^{2}P}{\sigma^{2}}}{{where}\mspace{14mu}\sigma^{2}} = {\frac{N_{0}}{2} \cdot {\frac{2}{T}.}}} & (8)\end{matrix}$is the noise power.

The BER at the output of the RAKE receiver corresponding to the i^(th)antenna element 704 may be obtained from the knowledge of theprobability density function (PDF) of γ_(i). For example, if no codingis used and BPSK modulation is applied to the data sequence inaccordance with the methodology described, for example, in “DigitalCommunications”, J. G. Proakis, 3^(rd) Edition, McGraw-Hill Series,1995, the BER is found by integrating the conditional error probabilityrepresented by Q(√{square root over (2γ_(i))}) over the PDF of γ_(i),for example:BER_(i)=∫₀ ^(∞) Q(√{square root over (2γ_(i))})p _(γ)(γ_(i))dγ _(i)  (9.)

Once the BER is estimated for all receive antennas, the antenna element704 yielding the minimum BER is selected:

$\begin{matrix}{\min\limits_{{i = 1},{\ldots\mspace{14mu} n_{R}}}{\{ {BER}_{i} \}.}} & (10)\end{matrix}$where n_(R) represents the total number of receiver antenna elements.

As will be readily appreciated, as coding is added to the system (e.g.,turbo coding, convolution coding) and other modulation levels are used,the modeling function used in (9) to estimate the BER will need tochange. Some embodiments according to some aspects of the presentinvention may provide that any fitting function which accurately modelsBER behavior for a given system may be used by an exemplary antennaselection algorithm. The fitting function will generally be dependentupon parameters including, for example, one or more of the following:the channel, coding and modulation used, signal processing at transmitand/or receiver side, receiver SNR and other parameters.

The exemplary embodiment illustrated in FIG. 7 may be extended to atwo-dimensional RAKE receiver in which processing is conducted in boththe space and time domain. In this context, an exemplary antennaselection algorithm may be incorporated to select a subset of N antennas(N>1), from a total of M antennas (M>N), which minimize the BER at theoutput of the 2D-RAKE.

While the present invention has been described with reference to certainembodiments, it will be understood by those skilled in the art thatvarious changes may be made and equivalents may be substituted withoutdeparting from the scope of the present invention. In addition, manymodifications may be made to adapt a particular situation or material tothe teachings of the present invention without departing from its scope.Therefore, it is intended that the present invention not be limited tothe particular embodiments disclosed, but that the present inventionwill include all embodiments falling within the scope of the appendedclaims.

1. In a transmitter having one or more RF chains and a plurality oftransmit antennas capable of transmitting RF signal energy to areceiver, an antenna selection method comprising: connecting the one ormore RF chains to an initial subset of the plurality of transmitantennas; determining a bit error rate of the receiver associated witheach transmit antenna within the initial subset of the plurality oftransmit antenna; comparing each bit error rate to a predefinedthreshold; disconnecting, when one the bit error rate exceeds thepredefined threshold, an associated transmit antenna within the initialsubset of the plurality of transmit antennas from a first of the one ormore RF chains; and connecting one of the plurality of transmit antennasto the first of the one or more RF chains, the one of the plurality oftransmit antennas not being included within the initial subset of theplurality of transmit antennas.
 2. The method according to claim 1,wherein the plurality of transmit antennas are greater in number thanthe one or more RF chains, and wherein a size of the initial subset ofthe plurality of transmit antennas is equivalent to a number of the oneor more RF chains.
 3. The method according to claim 1, wherein thetransmitted RF signal energy comprises at least one of: a code divisionmultiple access signal, a single carrier signal, an orthogonal frequencydivision multiplexed signal and a UWB signal.
 4. The method according toclaim 1, wherein the transmitted RF signal energy comprises a UWBsignal.
 5. The method according to claim 1, wherein the transmitted RFsignal energy comprises a spread spectrum signal.
 6. The methodaccording to claim 1, wherein the plurality of transmit antennas is fourtransmit antennas, wherein the one or more RF chains are two RF chains.7. The method according to claim 1, wherein the transmitter and thereceiver are part of a multiple-input-multiple-output (MIMO) system. 8.The method according to claim 1, comprising: demultiplexing an inputsignal into N independent substreams, wherein N is an integer greaterthan one; sending each of the N independent substreams along arespective path through N RF chains, wherein the one or more RF chainsare the RF chains; and converting, in N digital-to-analog converters,the N independent substreams into N analog independent substreams, eachof the N RF chains comprising one of the N digital-to-analog converters.9. The method according to claim 8, comprising: upconverting, in Nmixers, the N analog independent subsreams, each of the N RF chainscomprising one of the N mixers.
 10. The method according to claim 9,comprising: sending the N upconverted analog independent substreamsthrough N switches, each of the N RF chains comprising one of the Nswitches, wherein a respective switch of a corresponding RF chain beingconfigured to connect and to disconnect the corresponding RF chain andone of the plurality of transmit antennas.
 11. The method according toclaim 1, comprising: determining the initial subset of the plurality oftransmit antennas that are connected, via switches that are controlledby a switch controller, to the one or more RF chains.
 12. The methodaccording to claim 11, wherein the switch controller has informationthat identifies the initial subset of the plurality of transmitantennas.
 13. The method according to claim 1, wherein the transmitterand the receiver are part of aspatial-multiplexing-multiple-input-multiple-output-orthogonalfrequency-divisional-multiplexing system.
 14. The method according toclaim 1, comprising: weighting, by a set of complex coefficients, andcombining a single stream of symbols to generate a set of N outputsignals, wherein the one or more RF chains are N RF chains, wherein N isan integer greater than two.
 15. The method according to claim 14,comprising: sending the set of N output signals through the N RF chainsto produce N RF signals.
 16. The method according to claim 15,comprising: coupling the N RF signals to N antennas, via switchingcircuitry, to N of the plurality of transmit antennas, wherein a numberof transmit antennas is greater than N.
 17. The method according toclaim 15, comprising: transmitting the N RF signals through a singlechannel.
 18. The method according to claim 15, wherein the transmitterand the receiver are part of asingle-channel-multiple-input-multiple-output system.
 19. The methodaccording to claim 15, wherein the transmitter and the receiver are partof asingle-channel-multiple-input-multiple-output-orthognal-frequency-division-multiplexingsystem.